Frequency normalization in speech sound waves



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United States Patent 3,052,757 FREQUENCY N ORMALIZATION IN SPEECH SOUNDWAVES Meguer V. Kalfaian, 962 Hyperion Ave., Los Angeles, Calif. FiledDec. 17, 1958, Ser. No. 781,103 1 Claim. (Cl. 179-1) This inventionrelates to normalization of basic resonances in speech sound waves, andis particularly an improvement over the systems disclosed in my US.Patents No. 2,705,260 March 29, 1955; No. 2,708,688 May 17, 1955; andpatent application Serial No. 723,510 March 24, 1958 now Patent2,921,133, Ian. 12, 1960. Its main object is to provide improved methodsand means for standardizing the frequency positions of the basicresonances of the propagated speech sound waves, prior to analysis, forfinal translation into visible intelligible indicia, for example, byelectric typing devices.

In order that a machine, or the like, may be devised to simulate theinterpretive mechanism of human intelligence, in printing spoken words,as spoken" by all qualities and ranges of voices, without environmentalcontrol adjustments, or pre-adjustments to any particular voice, it isnecessary that all environmental variables are first standardized duringpropagation of the sound Waves, so that standard sets of parameters maybe derived to collectively define different phonetic sounds of thespoken words. To accomplish such standardization, advantage is taken ofthe fact that all phonetic sounds are composed of definite sets ofresonances whose ratios in frequency positions with respect to afundamental remain constant, no matter what band of the voice spectrumthey are produced in; this theoretical concept is disclosed in my abovementioned patents and applications. Accordingly, the object of thepresent invention is to select the varying fundamental frequenciesduring propagation of the sound waves, and shift all frequencycomponents to regions where their frequency ratios become constant withrespect to a pre-assigned fundamental frequency. The said sets ofparameters may then be derived, without variables, to collectivelydefine each phonetic sound of the spoken words in speech.

Frequency standardizing methods and systems had been disclosed in mypatent No. 2,708,688 May 17, 1955; and patent application Serial No.723,510 March 24, 1958. The systems utilized in both of thesedisclosures provide two storage tubes of the cathode ray type, in amanner that, the wave pattern containing in one cycle portion of theselected fundamental (during propagation of the sound waves) is recordedin one storage tube, and the wave pattern containing in the followingone cycle portion of the selected'fundamental is recorded in the otherstorage tube. While the first recording is processed, its time length(from inception to termination of the wave pattern) is measured andstored in the form of a first signal quantity.

Then, while the second recording is processed, the first recorded wavepattern is reproduced under control of the first quantity, so adjustedthat, the first recorded Wave pattern is reproduced in a predeterminedstandard time base period. The same process is repeated with the secondrecorded wave pattern, so that the end result is a cyclic reproductionof the wave patterns of the propagated sound wave at a standard timebase period. In order to allow time for reproduction of the recordedwave patterns prior to the arrival of successive wave patterns, thestandard time base period is adjusted to be several times shorter thanthe shortest time base period occurring in ordinary speech sound waves.Thus, the number of reproduced wave patterns will be many more (randomlyvarying) than the actual recorded wave patterns, which condition isfound advantageous for more accurate analysis of the ice wave patterns.The storage tubes presently available, however, are not ideally suitablefor the present purpose, as far as performance is concerned, andaccordingly, improved systems are disclosed herein to provide thenecessary functional performance. These systems will be described in thefollowing specification, with reference to the drawings, wherein:

FIG. 1 is a block diagram of the frequency transposing system.

FIG. 2 is a schematic diagram of the scanning system for frequencytransposition according to the invention; and FIG. 3 are waveformsinvolved in describing the arrangement of FIG. 2.

FIG. 4 is a schematic diagram of a sample-storage distribution system,in conjunction with the scanning system of FIG. 2, utilizing beamswitching tubes.

FIG. 5 is a detailed schematic diagram of the storage system inconnection with the circuitry of FIG. 4.

FIG. 6 is a modification of FIG. 4, utilizing transistors.

In describing first the broader aspects of the invention, the blockdiagram of FIG. 1 shows how the complex frequency components of theoriginal propagated sound Waves are transposed to standard frequencylocations utilizing memory tubes of conventional types. The speech soundwaves originating in block 1 are first applied to the block offundamental frequency selector 2 and the memory tubes, as represented bythe blocks 3 and 4. The function of the fundamental frequency selector 2is to produce at its output pulse-signals coincident with the arrivalsof wave trains from block 1; a wave train is one cycle portion of thefundamental frequency of the speech sound waves. These pulse-signals areapplied to an alternate switch, as represented by block 5, whichalternates its state of operation at each arriving signal-pulse, andimparts the operation of a two section scanning system consisting ofblocks scan-write 6, scan-read 7; and scan-read 8, scan-write 9,alternately in corresponding time periods with the arriving wavepatterns from block 1 (the expressions of write and read willhereinafter be referred to as recording and reproducing a wave signal,respectively). Each section of the scanning system is arranged tooperate an associate memory tube, for example, the section consisting ofblocks 6, 8 operates the memory tube in block 3, and the sectionconsisting of blocks 7, 9 operates the memory tube in block 4. Theoperational sequency of memory tubes 3 and 4 is such that, during activeperiod of scan-write block 6 the speech sound wave from block 1 isrecorded in the memory tube of block 3, while during this same timeperiod the scanread block 7 becomes active and a previously recordedwave pattern in memory tube of block 4 is reproduced and amplified inthe amplifier block 10. This particular performance is alternated when asucceeding signal-pulse from block 2 alternates the state of operationof switch block 5, e.g., scan-write block 6 and scan-read block 7 becomeidle, and scan-read block 8 and scan-write block 9 become active;effecting reproduction of the recorded wave pattern in memory tube ofblock 3, and recording of a new wave pattern in memory tube of block 4.

During recording periods of either one of the memory tubes 3 or 4, theassociated scan-write blocks, for example, block 6 or block 9, producesa scanning time base wave (saw tooth wave) of first constant timeperiod. Thus, the recording proceeds at a normal time base period.During reproduction time periods, however, the scan-read blocks 7 and 8produce scanning time base waves (saw tooth waves) of second constanttime periods having much shorter time than the first. The amplitude ofthe second scanning time base waves is varied stepwise in accordancewith the amplitude of the recording time base waves, so that eachrecorded wave in any one of the memory tubes of blocks 3 and 4 isreproduced in full scale. Thus, each recorded wave pattern is reproducedat a constant time period, no matter what its original recording timehad been. Due to the very short time chosen as a standard reproductiontime base, each recorded wave pattern may be reproduced several times,(the number of reproduction depending variably upon the ratio ofrecording and reproducing periods) before reproducing the succeedingwave pattern. With such standardized time base reproduction of theoriginal wave patterns of the sound waves from block 1, all basicresonances collectively representative of phonetic sounds are shifted toregions where their frequency ratios remain constant with respect to astandard fundamental frequency, a onecycle period of which isrepresented by said second time base period. The common output wave ofmemory tubes in blocks 3 and 4 is then amplified by the block 10, foranalyzing and deriving therefrom sets of resonances which collectivelydefine the phonetic characters of the spoken words in speech.

The memory tubes of blocks 3 and 4, as described in my above mentionedpatents and patent applications, had been mentioned to be of theexisting storage tube types. These storage tubes vary in types, and eachtype has its disadvantage functionally adaptable to the particularpurpose involved herein. The requirements for a satisfactory functionaloperation are, first, sufiicient number of storage elements forrecording a wave pattern during a line scan; reproducing the recordedwave pattern several times without deteriorating its original waveform;elimination of write to read switching pulses entering the reproducedsignal; and elimination of a sequence of erasing and priming functionprior to recording action. The storage systems described in thefollowing specification are, accordingly, contemplated to provide theserequirements.

Instead of utilizing elemental areas of a storage surface for recordinga signal wave, such as used in storage tubes of the cathode ray type, itis contemplated herein to use a large number of capacitors which arecharged sequentially, during recording time period, in sampledquantities proportional to the varying amplitudes of a wave pattern tobe recorded. These stored quantities are then resampled sequentiallyduring reproduction time period, so as to reconstruct the original wavepattern. Due to the large values that may be chosen for said capacitors,resampling of the stored quantities may be made several times withoutaffecting their original storage. The values of these capacitors,however, are small enough to be discharged without appreciable loss oftime for storage of new signal quantities.

To simulate the elemental sampling function of a cathode ray storagetube during a scanning line period, a pulse generator is first renderedto produce sampling pulses at a normal frequency rate, during writescanning time base period, and its frequency is shifted abruptly duringread scanning time base period to a position where the same number ofsampling pulses are produced during a standard time base period.

Production Write and Read Sampling Pulses The frequency of an oscillatormay be varied by an applied modulating voltage, for example, by varyingthe grid bias of a multivibrator. Such oscillators, however, are alsosensitive to plate voltage supply variations, and accordingly, requiresevere automatic control adjustments. The circuit arrangements disclosedherein are intended to operate satisfactorily with less severeadjustments.

The schematic arrangement of FIG. 2 will provide production of pulses atwidely varying frequencies, at the outputs of two independent branches,designated as OP-I and OP-Ia. The arrangement will also provideproduction of pulses at standard time base intervals, at the twoindependent outputs OP-II and OP-IIa. The two branches operatealternately, in a manner that, during one selected wave-pattern of thespeech sound wave the output OP-I will produce pulses at a normalfrequency rate for sampling and storing said wave-pattern, and theoutput of OP-II will remain idle, while simultaneously, the output OP-Iawill produce pulses at a shifted frequency rate for resampling andreproducing previously stored samples, between the standard intervals ofpulses produced at output OP-IIa. Since the operation of both of thesebranches is identical, with the exception of their operating timesequence, reference will mostly be made to the branch shown in the uppersection of the drawing.

The variable frequency oscillator consists of a saw-tooth wave generatorcomprising capacitor C1 charging in series with resistor R1 to thepotential of battery B1, and a discharger VI of capacitor C1. The RCtime constant of capacitor C1 and resistor R1 is adjusted to a normalfrequency for sample-storage during recording time periods. Inoperation, the capacitor C1 starts charging in series with resistor R1to the potential of B1, at a charging rate depending upon said RC timeconstant. A flyback trigger time is determined by a fixed potentialdeveloped across the cathode circuit resistor R2 of conducting tube V2.As the voltage across charging capacitor C1 equals the potential acrossR2, current starts flowing through resistor R3 in series with diode D1.The negative voltage developed at anode terminal of diode D1 is appliedupon the control grid of amplifier tube V3, through coupling capacitorC2, and further applied upon the control grid of normally inoperativedischarger tube V1, in positive polarity by way of phase inversionacross plate circuit resistor R4 and coupling capacitor C3. When thedischarger tube V1 approaches its threshold of conduction by thearriving positive voltage upon its control grid, the capacitor C1 startsdischarging, the action of which feeds back a regenerative positivevoltage upon the control grid of discharger tube V1. As thisregeneration increases, the current passing through V1 increases with anincrease in discharge speed of the capacitor C1. High speed discharge ofcapacitor C1 is also achieved by the fact that the anode current of V1passes through high voltage battery B2; rendering V1 highly conductiveduring the entire discharge period of capacitor C1. Due to the platesupply potential of B2, the current passing through V1 will startcharging capacitor C1 in reverse polarity immediately, after dischargeof C1 is completed. To avoid recharge of C1, a diode D2 is included,which offers low impedance and assume all the current that wouldnormally pass through C1 during its recharge period; thus effecting fastdischarge of C1, but avoiding recharge of same. In order to hastenrecovery of the non-conductive state of discharger tube V1 immediatelyafter discharge of C1 has been completed, the coupling capacitor C2 mustbe chosen of small value, and accordingly the value of load resistor R5must be chosen low enough to complete the discharge of couplingcapacitor C2 in a negligible time period, right after the discharge ofC1 has been completed. The value of coupling capacitor C2 must also bemuch smaller than the value of capacitor C1, so as to avoid loading uponthe latter. Thus, it is seen that a saw-tooth wave generation isachieved across capacitor C1 with very fast retrace time periods. Byadjusting the values of resistor R1 and capacitor C1 to have a fixed RCtime constant, the frequency of saw-tooth wave oscillation may then bechanged substantially linearly by varying the potential across R2.

As mentioned in the foregoing, a normal frequency of sampling pulses arerequired to be produced during recording time of an incoming wavepattern of the speech sound waves in bits of storage samples. Thesepulses may be derived from the flyback voltages of the saw-tooth wavesfrom across capacitor C1, by a small dilferentiating capacitor 04 at thecathode terminal of cathode follower tube V4. This tube is used as abufier stage, so as to avoid loading effect upon the charging capacitorC1; and it may be eliminated, if so desired. The output may also betaken from the plate circuit resistor R6, instead of the cathode circuitresistor R7. During reproduction period of above said stored samples,the frequency of saw-tooth wave oscillation is shifted for resamplingthe stored samples in a standard time period. As stated in theforegoing, the frequency of saw-tooth wave oscillation is shifted byshifting the voltage across R2. Also, this frequency shift must beabrupt at the end of recording time period, and remain in steady stateduring reproduction period. These two different voltages may be isolatedfrom each other, as in the following:

During recording of a wave pattern in bits of samples, the frequencyadjusting voltage is derived from across resistor R2 by way of thesteady state current passing through it in series with tube V2. Duringreproduction period, however, this frequency-shifting voltage is derivedfrom across resistor R8 in series with diode D3. In order to isolate thevoltages across R2 and R8 acting simultaneously, for frequencyadjustment of the oscillator, the voltage across R8 is raised above thevoltage across R2, during recording time period, by conduction of tubeV5. By raising the voltage across R8 above the voltage across R2,flyback triggering occurs at the peak of voltage across R2, and therebythe voltage across R8 becomes isolated from the voltage across R2. Theon-and-oif conduction of this tube is controlled by V6, which drawscurrent through resistor R9 during reproduction period, causing a highcurrent-cut-off negative potential upon the control grid of tube V5, andrelease this cut-off potential during recording time period bynon-conduction of tube V6, according to switching voltages applied uponthe control grid of tube V6. These switching voltages are graphicallyillustrated adjacent to tube V6, as indicated by their proper polaritiesduring write and read time periods. These switching voltages are alsoshown graphically in FIG. 3, in conjunction with various other switchingvoltages at diiferent phases, and reference to their application willhereinafter be made by the designated letters, for example, the wave atB in FIG. 3 is applied upon the control grid of tube V6 at terminal (B)in FIG. 2.

During production of saw-tooth waves at normal frequency, the controlgrid of tube V7 receives at terminal (A) a positive voltage (theswitching voltage A in FIG. 3), which drives it to conduction and drawscurrent through resistor R10. The negative voltage developed across R10drives tube V8 to non-conduction, which in turn releases a positivepotential that it had previously developed across resistor R11 byconduction therethrough. The cathode terminal of diode D4 now seesnegative potential, and allows current to flow through resistor H11 inseries with capacitor C5, to the fixed potential developed across R12.While this latter potential may be obtained from a fixed division acrosspower supply source B1, it has been shown developed across R12 by thecurrent passing in series with cathode follower tube V9, the controlgrid of which receives a fixed potential from across B1 by the voltagedividing resistors R13 and R14. The purpose of V9 is to act as a voltagecontrol tube, and keep the voltage across R12 constant during thecharging of capacitor 05, which evidently will load the supply voltage.The bypass capacitor C6 will help to keep the supply potential acrossR12 constant, if the control tube V9 alone is not sutficient for thepurpose. Of course, this fixed potential may be obtained otherwise, ifso desired.

Unlike linear saw-tooth wave generation, the RC time constant ofcapacitor C5 and resistor R11 is adjusted to render the non-linearcharging curvature useful, e.g., the capacitor C5 is allowed to chargenear the maximum of potential across R12, during the longest time periodoccurring in a wave pattern of the speech sound waves. This is becausesaw-tooth wave generation by capacitor C1 in series with resistor R1will change substantially linearly by linear voltage variation of thevoltage across resistor R8; and square-root variation is required forfrequency multiplication during reproduction time base periods. Thus thevoltage change across capacitor C5 must follow a square-law curvaturewith time, so that the voltage reduction across R8 may correspondinglyfollow a square-root curvature. In operation, the capacity C5 startscharging at the instant the tube V8 becomes non-conductive, until thelatter tube becomes conductive again by a negative switching voltageapplied at terminal (A). At this point, a large positive potential isdeveloped across resistor R11, and the capacitor C5 retains itsterminated potential thereon. Simultaneously, a positive potential isapplied upon the control grid of tube V6, at terminal (B); rendering V5non-conductive. The negative potential across capacitor C5 is appliedupon the control grid of tube V10 with proportional voltage reductionacross cathode circuit resistor R8. Since now the voltage across R8 islower than the voltage across R2, the triggering action takes place bythe trigger current passing through R3 and diode D3. Since also, thistriggering action takes place sooner than the charging voltage acrosscapacitor C1 reaching its normally designated level, the frequency ofsaw-tooth generation increases. Ideal accuracy of frequencymultiplication is not achieved by this system, but approximation willsuflice; as the high ratio between sampling frequencies and time basefrequencies will reduce the percentage of objectional errors. Theseerrors, however, will be substantially stabilized, and associatedcircuitry may be devised accordingly. Stabilization is obtained due tothe fact that, voltage variation of the supply battery B1 will effectproportional variations at all points of sensitivity, as long as the RCtime constants of C1, R1, and C5, R11 remain constant. For example, thecritical adjustments are to keep the sensitivity responses of tubes V2and V10 constant with variation of supply voltage source B1. Sincehowever, no amplification is involved, stabilization of cathodefollowers are simpler to retain under severe variations of supplyvoltage.

Generation of Pulses at Standard Time Base Intervals When the chargingof capacitor C5 stops by conduction of tube V8, the saw-tooth wavegeneration across C1 is shifted in frequency according to the voltagestored in capacitor C5. At this instant, reproduction of pulses atstandard time base intervals commences. This is achieved by a secondsaw-tooth wave oscillator comprising capacifor C7, which charges inseries with diode D5 and resistor R15. The discharge of capacitor C5 isaccomplished by normally inoperative discharger tube V11. Also, theonand-off switching of these oscillations is accomplished by tube V12,which under conduction draws current through resistor R15, and stopscharging of the capacitor C7 by a high negative voltage developed acrossresistor R15.

At the beginning of standard time base wave production, it is desirablethat the capacitor C7 is completely discharged. This is done by applyinga positive pulse upon the control grid of discharger tube V11, whichbecomes conductive and draws current through capacitor C7 and batteryB2; discharging C7. As described previously, a hold-back diode D6 isincluded; preventing recharge of C7 in reverse direction. The positivepulse upon the control grid of discharger tube V11 is applied atterminal (C), in series with resistor R16, by the pulses at C in FIG. 3.Also, the switching on-and-off voltage upon the control grid of tube V12is applied at terminal (A), by the square waves at A in FIG. 3.

When the control grid of tube V12 is driven highly negative, and V12becomes inoperative, the capacitor C7 starts charging in series withdiode D5 and resistor R15, across battery B1. As the voltage across C7reaches the voltage level at the junction terminal between voltagedividing resistors R17 and R18, current passes through resistor R19 anddiode D7, and consequently a negative voltage drop across resistor R19is transmitted to the control grid of amplified tube V13 throughcoupling capacitor C8. This negative voltage is amplified across platecircuit resistor R20 of the tube V13, in positive polarity, and furtherapplied upon the control grid of discharger tube V11 from across loadresistor R21. As the applied positive voltage upon the control grid oftube V11 reaches a threshold of anode conduction, the capacitor C7starts discharging; the action of which circulates a regenerativepositive voltage upon the control grid of discharger tube V11 withhighly speeded discharge of capacitor C7. After discharge of capacitorC7 is completed, the discharger tube V11 becomes non-conductive, asdescribed in the foregoing, and recharge of the capacitor C7 repeats;for continuous generation of saw-tooth waves at standard time baseperiods. The required pulses are derived from the flyback voltages bydifferentiating output capacitor C10, at the cathode circuit of V14. Theoutput OP-II may be taken either from the anode circuit resistor R22, orthe cathode circuit resistor R23. The frequency of these time base wavesis determined by the RC time constant of capacitor C7 and resistor R15,and also the voltage at the junction point of voltage dividing resistorsR17 and R18. As the voltage ratio at the junction point of resistors R17and R18 remains constant, regardless of voltage variations of B1, thefrequency of these standard time base waves can be kept constant bykeeping the values of capacitor C7 and resistor R15 constant.

When generation of time base waves across capacitor C7 is ended, byconduction of V12, a highly negative pulse voltage is applied upon thecontrol grid of tube V15, at terminal (D), by the waves at D in FIG. 3.As tube V15 becomes non-conductive by said negative pulse, the normalplate current passing through resistor R24 is released, and tube V16becomes conductive (from normal non-conductive state) to dischargecapacitor C; for a new start. The diode D8 is used to avoid reversecharge of the capacitor C5, as described in the foregoing. It will benoted that a large current will pass through R12 during conduction oftube V16, which would disturb the steady state voltage across thisresistor, were it not for the regulation by control tube V9. Of course,closer regulation may be obtained by more intricate circuitry, aspracticed conventionally, and may be utilized if so desired.

As explained in the foregoing, pulse generation by the arrangement inFIG. 2 is performed in two separate replica branches in alternatesequence of the incoming wave patterns of the speech sound waves. Inorder to show the sequence of these operations, the input switchingwaves, as shown graphically in FIG. 3, are designated by the properletters, as applied to the proper terminations in both branches of FIG.2. Also, due to the similarity in circuitry of both branches, numeraldesignations to component parts are included only in one branch, asshown.

Pulse Distributor Utilizing Beam Switching Tubes FIG. 4 is anarrangement of sample storage distribution system, responsive to theoutput pulses at OPI and OP- II in FIG. 2. Only one branch of storagedistribution arrangement is shown in FIG. 4, as the alternate branchwould be identical, and operated by the output pulses at OPIa and OPIIain FIG. 2. The circuit arrangement utilizes magnetron beam switchingtubes BST-l to BST-S, which in combination provide 360 switchingoperations for independent storage of signal samples. Each one of thesetubes contains ten switching targets, the first four of which make useof only nine targets in each tube, and the fifth one makes use of allten targets contained therein.

Beam switching tubes are used in the practice of electroncs, forexample, the type 6700 as manufactured by Burroughs Corp. For a detailedspecification of magnetron beam switching tubes, reference may be madeto an article, entitled New Applications for Beam Switching TubesElectronics, pp. 122-126, April 1956. As a brief reminder of itsfunction, however, each beam switching tube, for example, BST-l in FIG.4, contains ten targets (only three targets are shown in the drawing, inhorizontal lines), and ten spades (only three spades are shown in thedrawing, in 30 angles). There are also included ten odd and ten evengrids for controlling the position of the beam. All odd and even gridsare internally connected in parallel, respectively. In some tubes, aseparate terminal from one of the even grids is brought out of thevacuum envelope for zero positioning of the beam. This singular terminalis designated as G1; the parallel-connected even grids is designated asG2; and the parallel-connected odd grids is designated as G3. Inoperation, the beam projection may be formed upon any one of the targetsby first lowering the potential of its associate spade. For example,referring to BST-l, the beam may be initially formed upon target numberone, by lowering the spade potential across spade circuit reistor R25.Once the beam is formed on any target, it will flow in steady state, andmay be shifted sequentially from one target to another by alternatenegative pulses applied upon the odd and even grids. With reference tothe example of BST-l, alternate negative pulses upon the odd and evengrids (G2 and G3) will not shift the beam from target number one to theadjacent target, until a negative pulse is applied to the separatedeven-grid G1. By this brief description of the function of magnetronbeam switching tubes, operation of the circuit arrangement in -FIG. 4may be described as follows:

The output pulses at OP-I in FIG. 1 are applied to the input of aflip-flop trigger circuit 11 (shown in block diagram), so as to produceat its output alternate negative pulses, to be applied upon the odd andeven control grids of EST-1 through EST-4 in parallel, from across loadresistors R26 and R27. Diodes D9 and D10 are connected in parallel withresistors R26 and R27, respectively, and so polarized that, theysuppress any positive potentials developed across said resistors. Theoutput negative pulses at OPII, in FIG. 2, are applied upon the controlgrid of norm-ally conducting tube V17 (in FIG. 4), which becomesnon-conductive momentarily and releases the currents passing throughplate circuit resistors R28 to R31. The release of forward currentpassing through resistor R28 from base to emitter elements of transistorQ11 drives it to non-conductive state, and cuts off the plate supplypotential directly from spade electrodes of EST-1 to EST-5, and also theplate supply potential from target electrodes of BST-l to BST- 5 inseries with transistors Q1 to Q10. During this sh rt pulse period ofplate supply potential isolation, the switching tubes BST-l to EST-5 arecleared from their operating states. Simultaneously, a positive pulsefrom the anode circuit terminal of V17 is applied upon the control gridof normally inoperative tube V18, through coupling capacitor C11 andfrom across load resistor R32. The load resistor R32 is bypassed by asmall capacitor C12, so that the positive pulse applied upon the controlgrid of tube V18 subsides slightly later than the applied pulse uponcontrol grid of V17. Thus when BST-l to BST-S are cleared from theiroperating states, the delayed conduction of tube V18 passes currentthrough spade circuit resistors: R25 of BST-l; R33 of EST-2; R34 ofEST-3; R35 of BST-4; and R36 of BST5; in series with isolating diodesD11 to D15, respectively, and by lowering the voltages upon thesespades, the beams of these switching tubes are reset upon these spades,and consequently upon their associate targets, for example, targetnumber one of BST-l; targets number zero of BST-2 to EST-4; and targetnumber one of BST-S. Also simultaneously, a positive pulse from theanode circuit terminal of V17 is transmitted to the flip-flop block 12,through coupling capacitor C13, for resetting the flip-flop circuit ofblock 12 to a position as to produce a first output negative pulseacross load resistor R37 when an operating pulse arrives at its inputterminal 13; for exciting the even grids of BST-S. If the flip-flopcircuit of block 12 is designed to respond to a negative pulse forresetting, then the positive pulse arriving from anode circuit of tubeV17 is first phase inverted by a conventional mode.

With the above referenced initial settings of BST-l 9 to BST- (beamposition on targets one of BST-1 and BST-5; and beam position on targetsZero of BST-2 to BST-4), the gate transistor Q1 starts conducting by theforward current between base and emitter elements across resistor R38,through target electrode number one of the switching tube BST-5. Thiscurrent passes in series with the main gate-transistor Q11 and resistorR39, the reverse current through the latter of which cuts off conductionof the gate transistors Q2 to Q10. The arrival of input pulses atflip-flop trigger circuit in block 11 alternates its state ofconduction, and shifts the beam of BST-1 from target one to target ninesequentially. At the end of target nine when a following negative pulsearrives upon the even grids of BST-1, the beam shifts upon target zero,and remains locked in this position without further shifting; until anegative pulse is impressed upon the grid G1. During transition per odfrom target nine to target zero of BST-1, a positive pulse istransmitted to the control grid of amplifier tube V19, through couplingcapacitor C14. This positive pulse is amplified and phase inverted inthe anode circuit resistor R40 of V19, and applied upon the control gridG1 of BST-2, through coupling capacitor C15 from across load resistorR41. The beam of BST-2 shifts from target zero to target one, andthereon to target nine by the alternate negative pulses upon controlgrids G2 and G3. At the end of target nine, the following negative pulseupon even grids G2 shifts the beam upon target zero, and is locked inthis position. During this transition, a positive pulse is applied uponthe control grid of amplifier tube V20, through coupling capacitor C16.This negative pulse is amplified and phase inverted across plate circuitresistor R42, of V20, and applied upon the control grid G1 of BST3,through coupling capacitor C17, and from across load resistor R43;shifting the beam of BST-3 from target zero to target one. The followingnegative pulses upon control grids G2 and G3 shift the beam sequentiallyto target nine, and to target zero to get locked thereat. The positivepulse from target nine is applied upon the control grid of amplifiertube V21 through coupling capacitor C18. This positive pulse isamplified and phase inverted across plate circuit resistor R44, andapplied upon the control grid G1 of BST-4, through coupling capacitorC19 and from load resistor R45; shifting the beam from zero target toone target. The following alternate negative pulses upon control gridsG2 and G3 shift the beam sequentially to target nine, and to target zeroin a stationary position. During transition period from target nine totarget zero, a positive pulse is transmitted to the control grid of tubeV22, through coupling capacitor C20. This positive pulse is amplifiedand phase inverted across plate circuit resistor R46, and applied uponthe control grid G1 of BST-1, through coupling capacitor C21 and fromacross load resistor R47; shifting the beam of BST- 1 from zero targetto number one target. In this position of beam projections in BST1 toBST-4, the counting process continues in a cyclic mode. At this startingpoint, however, a pulse voltage is transmitted to the input of flip-flopcircuit in block 12, via coupling capacitor C22 and input terminal lead13. The flip-flop in block 12 alternates its state of conduction, andproduces a negative pulse across load resistor R37. This latter pulsevoltage is applied upon the even control grids of BST-5, shifting the"beam from target number one to target number two. At this point, theforward current through R38 is released, and a forward current is passedthrough resistor R48 by the number two target of BST-5. 'Ihus, gatetransistor Q1 is cut off from conduction, and current supply to the BSTswitching tubes is now supplied in series with gate transistor Q2 andthe main gate-transistor Q11. As the cyclic operation of BST-1 to BST-4continues, the beam of BST-5 keeps on shifting sequentially (once ateach said cycle) to the 10 tenth target, which in turn controls theon-and-oif operations of gate transistors Q1 to Q10.

Within one cyclic operation of beam switching tubes BST-1 to BST-4, 36useful operations are obtained, as one of the ten targets of these tubesis used as a holding position of the beam. The fifth beam switching tubeBST-5, is used as a gate control for each cyclic operation of BST-1 toBST-4. Accordingly, all ten targets of BST-5 are made useful, whichrenders 360 useful opera-tions. Each useful target of the BST-1 to BST-4draws current through ten gated resistors, in series with isolatingdiodes, and in series with separate gate-transistors. For example, thetarget number one of BST-1 draws current through resistor R1, in serieswith isolating diode D16, when gate-transistor Q1 is conducting. Also,this target one draws current through resistor R2, in series withisolating diode D17, when gate-transistor Q2 is conducting. Only tworesistors and two diodes.

are shown associated with each target; but actually ten resistors andten isolating diodes are used in conjunction with the transistors Q1 toQ10. Of course, any number of these parts may be utilized, according toa particular number of sample distribution desired.

The particular arrangement of resistors, for example, the resistors R28to R30, is used as one Way of dividing the forward current from base toemitter of transistors Q1 to Q11; as the high current imposed by V17 andthe targets of BST-5 might damage these transistors. 0f course, suchcircuitry can be modified, if so desired. Also, these transistors may bereplaced by vacuum tubes, if so desired. Further, the vacuum tubes V19to V22 may be replaced by transistors, if so desired. The diodes D9,D10, and D18 to D23 across load resistors R26, R27, R41, R43, R45, R47,R37 and R49, respectively, are used to suppress the positive pulsevoltages developed across said resistors; in application upon thecontrol grids of beam switching tubes BST-1 to BST-5. The beam switchingtubes BST-1 to BST-5 are shown having a common cathode bias circuitcomprising resistor R45 and bypass capacitor =C23. Due to characteristicvariation of these tubes in manufacture, however, separate cathode biascircuits may be used, if so desired. For the same reason, separate biassources may be used for the control grids of these beam switching tubes,instead of obtaining said bias from the common junction point of voltagedividing resistors R46 and R47 the latter being bypassed by capacitorC24.

Sample Storage System In reference to the arrangement of FIG. 4, theuseful sampling operations were referred to the pulse voltages developedacross plurality of resistors connected to each target of the beamswitching tubes BST-1 to BST-4, in series with isolating diodes. Theoutputs of these plurality of resistors are used as distribution signalsfor storage samples of the sound wave. The circuitry of this storagesystem is shown in detail in the arrangement of FIG. 5, wherein, thebeam switching tube BST may represent any one of the tubes BST-1 toBST-4 in FIG. 4. Accordingly, the tube BST in FIG. 5 will be referencedas a typical example of the operating conditions of any one of said beamswitching tubes.

Assume initially that a forward current is made to pass from base toemitter of transistor Q12, by a switching current from terminal (B).Assume also that the beam of BST (in FIG. 5) impinges upon target numberone, and current flows through resistor R48 in series with isolatingdiode D24. The negative voltage drop across this resistor is transferredand stored in storage capacitor C25, through coupling capacitor C26 andisolating diode D25, and also in series with the secondary coil oftransformer T1, transistor Q12, and bias battery B3. The bias battery B3is polarized in opposition to the voltage developed across R48, but isless in magnitude than the latter voltage, so that the storage incapacitor C25 is equal to the voltage developed across R48 minus thebias voltage of battery B3. During this storage period, a voltage(representative of the sound wave) of unpredictable polarity existsacross the secondary coil of transformer T1, which further determinesthe quantity to be stored in capacitor C25. For maximum operationalconditions, the bias voltage of battery B3 would be adjusted to equalhalf the voltage amplitude as developed across R48, and the peakamplitude of voltage developed across secondary coil of transformer T1equal the voltage of B3. With such adjustments, maximum modulation ofthe sample storage in capacitor C25, by the sound wave, will result.Thus When the beam of BST is projected upon target number one, andtarget current passes through resistor R48 in series with isolatingdiode D24, a sampled signal-quantity is stored in capacitor C25.Whereas, when said target current passes through resistor R49 in serieswith isolating diode D26, the pulse-voltage across R49 is transferred tostorage capacitor C27, through coupling capacitor C28; at a modulatedmagnitude. In a similar mode, when the beam of BST is projected upontarget number nine, and target current passes through resistor R50, inseries with isolating diode D27 the pulse-voltage developed across thisresistor is transferred to the storage capacitor C29, through couplingcapacitor C30; at a modulated magnitude. Also, when last said currentpasses through resistor R51 in series with isolating diode D28, thepulse-voltage developed across this resistor is transferred to thestorage capacitor C31, through coupling capacitor C32; at a modulatedmagnitude. Thus it is seen that, as the beam of switching tube BSTshifts from one target to another sequentially, modulatedsample-voltages are stored in capacitors C25, C27, C29, C31, et cetera,sequentially. In order to avoid cross modulation, the bias voltage byB3, and the peak modulating voltage should be less than the maximumallowed. The diodes D32 to D33 are used as load impedances at the openends of coupling capacitors C26, C28, C30, C32, etc.; and they may bereplaced by resistors, instead. The advantage of diodes, however, is tooffer high impedance during charging periods of the storage capacitors,and low impedance during quiescent periods, so as to recharge thecoupling capacitors at faster speeds.

When the forward current from base to emitter of transistor Q12 isswitched off from across resistor R52, at terminal (B), this transistorstops conducting, and the only electrical path is represented byresistor R53. As the beam of switching tube BST continues shifting fromone target to another, the storage capacitors are discharged in serieswith R53. But this resistor is chosen of high value, so that anydischarge during a transient period is negligible. Accordingly, theresistor R53 is coupled to the control grid of cathode follower tubeV23, so that the stored potentials across said storage capacitors areresampled and reproduced during the reproduction period for saidfrequency transposition. The original waveform of the sound is showngraphically at a, and the reproduced waveform is shown by the pulses atb, in proportional amplitudes. These output pulses may be obtainedeither from the anode circuit resistor R54 and coupling capacitor C33,or cathode circuit resistor R55 and coupling capacitor C34, of outputtube V23.

During rewrite period, the storage capacitors C25, C27, C29, C31, etc.;must be discharged quickly from their charged states, so that newsamples may be stored. The discharge of these capacitors is accomplishedthrough isolating diodes D34 to D37, in series with normally inoperativetransistor Q13. At the beginning of rewrite period, a forward currentpulse is applied to the base of transistor Q13, at terminal (D), whichrenders it conductive and discharges said storage capacitors in serieswith last said diodes. Thus, write and read conditions 12 are effectedby the circuit arrangement in FIG. 5, in sampled bits.

Due to characteristic variation of beam switching tubes in manufacturingprocess, it may be necessary that the voltage developed at the targetsof said tubes be clamped. This condition may be easily secured by addingdiodes D38, D39, etc., at these target terminals and connected inparallel to a reference voltage source. Since voltage isolation isrequired during clearance of beam switching tubes, however, thisreference voltage source must also be switched on and off at the propermoments. One way to secure this switching action is to produce thereference voltage in series with a grid controlled vacuum tube, so thatit may be switched on and off. As shown in the drawing, the referencevoltage is produced across cathode circuit resistor R56 of cathodefollower tube V24, the control grid of which is normally biased toonposition from across resistor R57 in series with normally operatingtransistor Q14. The base of this transistor is connected in parallelwith the base of transistor Q11 in FIG. 4, so that during reset periodQ14 is also rendered non-conductive, the condition of which impresses alarge negative voltage upon the control grid of V24, in series with B4,and thereby rendering V24 non-conductive during said reset pulse period.

Sample Distribution System, Utilizing Transistors FIG. 6 is anotherarrangement for sample distribution, wherein, transistor flip-flops areused, instead of beam switching tubes. The circuitry, however, isdifferent than conventional flip-flop counting systems, and it providessequential distribution of control pulses. The arrangement providespairs of cross-controlled flip-flop circuits, each having first andsecond outputs and driving inputs, the first and second inputs of whichare connected in parallel, respectively, and excited in alternate timeperiods.

The first of the first pair of flip-flop circuits comprises transistorsQ15, Q16, having collector circuit resistors R58, R59, and voltagedividing and coupling resistors R60, R61, and R62, R63. The second ofthe first pair of flip-flop circuits comprises transistors Q17, Q18,having collector circuit resistors R64, R65, and voltage dividing andcoupling resistors R66, R67, and R68, R69. The first of the second pairof flip-flop circuits comprises transistors Q19, Q20, having collectorcircuit resistors R70, R71, and voltage dividing and coupling resistorR72, R73, and R74, R75. The second of the second pair of flip-flopcircuits comprises transistors Q21, Q22, having collector circuitresistors R76, R77, and voltage dividing and coupling resistors R78,R79, and R80. Transistors Q15 to Q22 are chosen of the tetrode type, sothat collector current through any one of these transistors may be madeto depend upon simultaneous forward currents of the two independent baseelements contained therein, thereby providing two separate inputs foreach of said first and second driving inputs. These pairs of flip-flopsare arranged in sequential order, in a manner that, the said secondoutputs of succeeding flip-flops are directly coupled to one of said twoseparate inputs of said first inputs, and the first outputs ofreturn-order flip-flops are directly coupled to one of the two separateinputs of said second inputs. By such an arrangement, it is possible toshift the on-and-off operating states of succeeding flip-flopssequentially, by alternate driving signals, either in forward orbackward direction.

In operation, assume that transistors Q15, Q17, Q19, Q21 are initiallyconducting, and transistors Q16, Q18, Q20, Q22 are non-conducting. Whena negative pulse from flip-flop in block 14 is impressed upon resistorR82, backward current is imposed from base to emitter of Q15 and Q19;rendering these transistors non-conductive. Transistor Q16 of the firstflip-flop becomes conductive in a stabilized state. On the home returnof the pulse across R82, however, transistor Q19 becomes conductiveagain, due to the fact that, a forward current from one of its baseelements to emitter is constantly supplied by direct coupling from theoutput of conducting Q18, and a backward current from one of the baseelements to emitter of Q20 is constantly supplied by direct couplingfrom the output of non-conducting transistor Q21. Thus when a drivingpulse appears across R82, only the first flip-flop changes its state ofoperation. When the following negative pulse appears across resistorR83, forward current is imposed from one of the base elements to emitterelements of transistors Q17 and Q21; rendering these transistorsnon-conductive. When the input pulse across R83 subsides, the secondflip-flop, comprising Q17 and Q18, remains stabilized in the changedstate, whereas, the fourth flip-flop, comprising Q21 and Q22, returns toits original operating state, by reason of said direct intercouplings,as described by way of the first flip-flop. As these alternate negativepulses continue across R82 and R83, the flip-flops change their statesof conductions sequentially, until transistors Q15, Q17, Q19, Q21 arenonconducting, and transistors Q16, Q18, Q20, Q22 are conducting. Inthis condition, the direct intercouplings are in the reverse direction,and accordingly, backward sequence (fourth flip-flop toward firstflip-flop) of flip-flop operation may be effected by applying alternateinput negative pulses upon resistors R84 and R85, from the output offlip-flop circuit in block 15. For example, by applying a negative pulseacross R85, Q22 becomes non-conductive, and Q21 conductive. During thefollowing pulse across R84, Q20 becomes non-conductive, and Q19conductive; etc. Thus it is seen that the circuit arrangement of FIG. 6provides forward and backward counting, to increase the capacity ofcounting with lesser number of component parts.

The input driving pulses are received from block 16, through alternategate in block 17. This gate admits the pulses from block 16 to eitherblock 14 or block 15, according to its state of operation, which isdetermined and controlled by the output pulses of reset pulse producingblock 18.

The distributed signal pulses are stored in individual capacitors, asdescribed by way of the arrangement of FIG. 5, and only one typicalexample of storage is shown in the drawing. For example, the negativepulse produced across R58 is stored in storage capacitor C35, in serieswith diode D40, through coupling capacitor C36, and from across loadresistor R86. The output load resistor R87, and parallel connected diodeD41, represent the same component parts R53 and D29, in FIG. 5. Themodulating parts are not shown in FIG. 6, as these are already explainedpreviously. The diode D42 is shown to indicate that the storagecapacitor C35 is to be discharged through said diode, during re-writeperiod, as described previously, by way of FIG. 5. It will be well worthto mention here that the value of coupling capacitors C36 should bearranged to cause least cross-modulation, for example, when conductingtransistor Q19 is rendered nonconductive momentarily and on again; thedischarge period of said coupling capacitors not being fast enough tocause erroneous storage.

With the exemplary arrangements, as shown, it will be obvious to theskilled in the art that, various modifications, adaptations, andsubstitutions of parts may be made without departing from the spirit andscope of the invention.

What I claim is:

In speech sound waves containing a plurality of wave components inintegral harmonic relation to a fundamental component, but wherein thefrequency of said fundamental varies randomly, the system of shiftingthe resonances of said plurality of wave components to regions wheretheir integral harmonic relations remain constant with respect to areference fundamental component, the system comprising means forproducing speech sound waves having varying fundamental frequencycomponents; means for selecting and forming alternate switching waves atthe varying fundamental frequencies from said produced waves; an onandoff-gating means having input and output; means for applying theproduced sound waves to the input of said gating means, and means foroperating the gating means in on-and-off positions by said switchingwaves, thereby producing the sound waves in on-and-off states at theoutput of said gating means at said selected frequencies; afrequency-controlled pulse wave generator normally adjusted forproducing pulse waves at a sampling frequency of the produced soundwaves; a pulsed sample storage means; means for sampling the outputwaves of said gating means by said pulse waves, and means for storingthese samples in said storage means; a reproducing means for reproducingsaid stored samples under control of pulse waves; a counting means forrecording the number of said samples stored in the storage means; anormally inoperative first coupling means between the counting means andsaid generator; a normally inoperative second coupling means betweensaid reproducing means and said generator; and means for applying saidalternate switching waves simultaneously to said first and secondcoupling means in a phase as to operate them in on-states when theproduced complex waves appear at the output of said gating means,whereby first, to impart reproduction of the stored samples by saidreproducing means under control of pulse waves from said generator, andsecond, said counted record to shift the pulse fre quency of saidgenerator by an amount as to produce approximately the same number ofpulses as said count during a prefixed reproducing time period equal toone wavelength period of the reference fundamental frequencyaforementioned, thereby effecting the desired frequency conversion ofthe original speech sound waves.

References Cited in the file of this patent UNITED STATES PATENTS2,705,742 Miller Apr. 5, 1955 2,708,688 Kalfaian May 17, 1955 2,921,133Kalfaian Jan. 12, 1960

